The fast, efficient and error-free transmission of digital information from one point to another has become increasingly important. Many communications systems exist which permit digital information to be transmitted over various types of communication channels, such as wireless channels, fiber-optic channels, and wire line channels.
The present invention relates to a receiver for DMT based (Discrete Multitone Modulation) signals and preferably for DMT ADSL (Asymmetric Digital Subscriber Line) and Lite ADSL modems. In particular, it relates to receiver windowing and allied techniques. Some related techniques have been previously considered as possible solutions to combat narrow band interference (NBI) and spectrally colored crosstalk for DMT transceivers. Windowing, or more generally, pulse/waveform shaping filtering, can be carried out over single or multiple DMT symbols and can be carried out jointly or singly at the transmitter and receiver. Several existing methods require the participation of the transmitter and/or utilize window/pulse shapes that span one or more DMT symbols and improve performance based on better spectral containment.
However, in the context of current ANSI and ITU-T DMT based ADSL standards, no pulse shaping is used at the transmitter. Thus, any methods which require the transmitter to perform waveform shaping will either have to be standardized or will result in a proprietary standard non-compliant modem. There exist methods which do not require transmitter participation but do require substantially equalized channels (e.g. NEC's ADSL modem has provision for partial shaping at the boundaries of the orthogonality interval).
The present invention will be described in the context of a wireline communications channel, such as a telephone line which utilizes a twisted pair of copper wires. It is noted that the use of the present invention is not limited to wireline systems as those skilled in the art will appreciate from the discussion hereinbelow.
A modem is typically used to transmit and receive digital data over a telephone line. Modems employ a modulator to transmit the digital data over the telephone line and a demodulator to receive digital data from the telephone line. One common modulation technique is known as discrete multi-tone modulation (DMT) which requires a discrete multi-tone transmitter and a discrete multi-tone receiver at each modem in a communication system. Often, those skilled in the art refer to such modems as employing a DMT physical layer modulation technique.
Reference is now made to FIG. 14 which is a block diagram of a conventional DMT communications system 1. The system 1 includes a DMT transmitter 10, a transmission channel 20, and a DMT receiver 30. The DMT transmitter 10 includes a symbol generator 12, an inverse fast Fourier transform (IFFT) modulator 14 and a cyclic prefix generator 16. The DMT transmitter 10 receives an input bit stream b(n) which is fed into the symbol generator 12. The symbol generator 12 produces a signal X(k) which is fed into the IFFT modulator 14. X(k) is a complex signal (i.e., a signal understood by those skilled in the art to comprise both a real and an imaginary component) formed by mapping groups of bits of the input bit stream b(n) into a complex data space such that the complex signal X(k) has a length of N samples. Symbol generator 12 also augments the signal X(k) with a complex conjugate to obtain a conjugate symmetric signal of 2N samples.
The IFFT modulator 14 performs a 2N-point inverse fast fourier transform on the conjugate complex signal X(k) to obtain the sampled real signal x(n). Since X(k) is a symmetric signal, the output of the IFFT modulator 14 is a real signal x(n). The real signal x(n) may be thought of as the summation of a plurality of cosine functions each having a finite length and a different frequency, phase, and amplitude, where these frequencies are multiples of a fundamental frequency. Since each of the cosine functions has a finite duration, x(n) is a varying amplitude discrete signal having a finite duration spanning 2N samples. Each cosine function is known as a bin or tone.
The transmission channel 20 is modeled as including a D/A converter 22, transmit filter (not shown), a receive filter (not shown), and an A/D converter 26 on either end of a wire loop 24. Those skilled in the art will appreciate that a practical system will employ the D/A converter 22 (and the transmit filter) in the DMT transmitter 10 and will employ the A/D converter 26 (and the receive filter) in the DMT receiver 30.
Those skilled in the art will appreciate that the frequency spectrum of x(n) may be thought of as a plurality of orthogonal (SIN X)/(X) functions, each centered at a respective one of the frequencies of the cosine functions of x(n).
x(n) is transmitted over the channel 20 to the DMT receiver 30. Since the transmission channel 20 has a non-ideal impulse response h(n), the received signal y(n) will not exactly match x(n). Instead, y(n) will be a function of the convolution of x(n) and h(n). Typically, h(n) will look substantially like the curve shown in FIG. 15. The non-ideal characteristic of h(n) introduces an amount of interference (specifically intersymbol and interchannel interference) which should be compensated for in both the DMT transmitter 10 and the DMT receiver 30.
A common technique in compensating for the non-ideal impulse response of the transmission channel 20 is to introduce a so-called guard band at the beginning of each finite duration signal x(n) to produce x′(n). The cyclic prefix generator 16 performs this function. The guard band is typically formed of the last G samples of x(n) for each DMT symbol. If the length of the impulse response h(n) of the transmission channel 20 is less than or equal to G+1, then the guard band of length G will be sufficient to eliminate the interference caused by the impulse response h(n). The guard band is commonly referred to in the art as a “cyclic prefix” (CP).
Unfortunately, the impulse response h(n) of a typical transmission channel 20 may be excessively long, requiring cyclic prefix lengths which substantially reduce the rate at which digital bits are transmitted across the transmission channel 20. The DMT receiver 30, therefore, employs signal processing techniques which effectively shorten the impulse response h(n) of the transmission channel 20, thereby permitting a corresponding reduction in the length of the cyclic prefix required at the DMT transmitter 10.
The DMT receiver 30 includes a time-domain equalizer (TEQ) 32, CP remover 34 for removing the cyclic prefix, a fast fourier transform (FFT) demodulator 36, and a bit generator 38. The time-domain equalizer 32 is a finite impulse response (FIR) filter designed to compensate for the non-ideal impulse response h(n) of the transmission channel 20. In particular, the time-domain equalizer 32 employs a finite number of coefficients (T) which are calculated to compensate for the non-ideal impulse response of the transmission channel 20. The time domain equalizer 32 operates on the impulse response h(n) of the channel 20 such that the combined impulse response heff(n) of the channel 20 and the time domain equalizer 32 has maximum energy within a limited band of samples. This may be thought of as “shortening” the effective impulse response of the channel 20. The output of the time domain equalizer is z′(n).
The CP remover 34 is employed to remove the cyclic prefix from z′(n) to obtain z(n). The signal z(n) is input into the FFT demodulator 36 (which is understood to include a one-tap per-bin frequency domain equalizer/AGC function) to produce the complex symmetric signal X(k). After the complex conjugate portion of the signal X(k) is removed, the bit generator 38 maps the complex signal X(k) into an output bit stream b(n), which theoretically matches the input bit stream b(n).
While the conventional DMT receiver 30 of FIG. 14 operates optimally when white noise is present at the output of the time domain equalizer 32, it is susceptible to increased interference when colored noise is present. This is particularly pronounced when the colored noise exhibits spectral nulls or spectral peaks.
Colored noise may be present at the output of the time domain equalizer 32 because (i) additive colored noise was injected into the signal x′(n) as it was transmitted over the transmission channel 20; and/or (ii) the time domain equalizer 32 itself introduces spectral shaping (especially spectral nulls/peaks) into the signal z′(n). Thus, even if the transmission channel 20 does not introduce additive colored noise into the received signal y(n), the time domain equalizer 32 may itself introduce spectral coloration into the additive noise of signal z′(n). Consequently, although the time domain equalizer 32 may produce a “shorter” effective impulse response heff(n), it may degrade system performance by introducing colored noise (especially spectral nulls/peaks) into z′(n). In particular, the rate at which data bits b(n) are transmitted over the transmission channel 20 and the error rate of such transmission may be adversely affected by colored noise at the output of the time domain equalizer 32.
Accordingly, there is a need in the art for an improved DMT communication system which is capable of (i) improved performance in the presence of noise, and in particular, narrowband interference and inter-bin interference; (ii) compensating for additive colored noise and/or narrowband interference (NBI) introduced by the transmission channel and/or other conditions leading to diminished orthogonality between bins including inadequate channel shortening, symbol timing offset and jitter; (iii) mitigating against the spectral coloration of additive noise by the time domain equalizer; (iv) suppressing side lobes caused by DFT frequency response; (v) obtaining better performance against cross-talk and narrowband interference (NBI) without changing the transmitter; and/or (vi) suppressing a specific type colored noise in the form of local echo signal cross-talk in the FDM duplexing method.